Low power signaling interface

ABSTRACT

In a chip-to-chip signaling system includes at least one signaling link coupled between first and second ICs, the first IC has an interface coupled to the signaling link and timed by a first interface timing signal. The second IC has an interface coupled to the signaling link and timed by a second interface timing signal that is mesochronous with respect to the first interface timing signal. The second IC further has phase adjustment circuitry that adjusts a phase of the second interface timing signal using a digital counter implemented with Josephson-junction circuit elements.

TECHNICAL FIELD

The present disclosure relates to high-speed signaling betweenintegrated circuit devices.

DRAWINGS

The various embodiments disclosed herein are illustrated by way ofexample, and not by way of limitation, in the figures of theaccompanying drawings and in which like reference numerals refer tosimilar elements and in which:

FIG. 1 illustrates a signaling system embodiment in which constituentintegrated circuit devices (IC's) signal one another from disparatetemperature domains and in which an overclocked timing reference is usedto enable purely digital signal timing adjustment;

FIGS. 2A and 2B illustrate exemplary receive and transmit timingcalibration operations that may be carried out to identify receive andtransmit boundary clock phases, respectively;

FIG. 3 illustrates a more detailed embodiment of asuperconducting-domain PHY (physical signaling interface) having receiveand transmit timing calibration circuitry implemented by rapidsingle-flux-quantum (RSFQ) circuit elements;

FIG. 4 illustrates an exemplary phase adjustment/selection operationthat may be carried out within any one of the receive clock generatorsshown in FIG. 3 to generate respective receive clock signals;

FIG. 5 illustrates a more detailed embodiment of a receive clockgenerator that may be used to implement any or all of receive clockgenerators within the RSFQ PHY of FIG. 3;

FIG. 6 illustrates embodiments of a self-timed modulo-16 phase counterand self-timed count-comparator, and also a self-timed count-matchregister that may store a 4-bit count-match value generated, forexample, by the clock-select counter of FIG. 5;

FIGS. 7A-7I and FIG. 8 illustrate operation andresistor-transistor-logic models of various RSFQ circuit elements shownin FIG. 6 and other embodiments described herein;

FIGS. 9 and 10 illustrate alternative phase counter embodiments havingshortened critical feedback paths between phase-count register outputand input to permit higher fast-clock (FCLK) frequencies andcorrespondingly higher signaling rates;

FIG. 11 illustrates exemplary per-bit count-match circuitry thatgenerates a set of match bits for the toggle counter slices shown inFIG. 10;

FIG. 12 illustrates an exemplary self-timing pulse generator that may beused to generate self-timing pulses (T1, T2, . . . ) in any of theself-timed RSFQ logic circuit embodiments herein;

FIG. 13 illustrates a plesiochronous receive/transmit timing approachthat permits a fast clock signal (FCLK) to have an arbitrary (includingnon-integer) frequency ratio with respect to a data-rate reference clocksignal (RefCK);

FIG. 14 illustrates an embodiment of a receive clock generator thatenables generation of a receive timing signal having a digitallycontrolled phase offset relative to RefCK despite a plesiochronousrelationship between FCLK and RefCK;

FIGS. 15, 16, 17 and 18 illustrate embodiments of digital phaseadjusters that avoid a reset-timing pinch by providing dual fast-clockgenerators and phase counters and activating an alternate fast-clockgenerator/phase counter pair from one RefCK period to the next; and

FIGS. 19 and 20 correspond to an alternative receive/transmit timingadjustment technique that lack encoded phase counters and (in at leastone embodiment) phase count comparators.

DETAILED DESCRIPTION

In various embodiments herein, signal receive/transmit operations areexecuted with respect to timing pulses selectively extracted from anover-clocked timing reference by digital timing-phase alignmentcircuitry. While the various disclosed embodiments may be deployed invirtually any environment and implemented using traditionalresistor-transistor logic (RTL) circuit elements, many of theover-clocked digital phase adjuster implementations are particularlywell-suited to deployment in cryogenic environments and implementationwith various superconducting devices (e.g., rapid single-flux-quantum(RSFQ) logic elements or other Josephson-junction-based logicfamilies)—devices in which conventional analog phase-adjust techniquesare difficult to implement or altogether impracticable. Moreover, insuch environments, it becomes possible to achieve extreme overclockingrelative to signaling data rate (e.g., clock rates ten to 100 or moretimes the signaling link bit rate) with very low power expenditure, evenwhere the counterpart signaling device (i.e., destination for outboundsignals and source of inbound signals) is disposed in a substantiallywarmer temperature domain. For example, in a number of embodimentsdisclosed herein RSFQ logic elements (or other Josephson-junctiondevices) are used to effect purely digital timing-phase adjustment byextracting timing events from an overclocked timing reference (“fastclock” or FCLK) while expending exceedingly low energy per bit, in somecases in the neighborhood of a pico-joule per bit or less.

In a number of mesochronous embodiments, the over-clocked timingreference is sub-divided by an over-clock factor N (e.g., N=5, 8, 10,16, 50, 100, etc.) to yield a data-rate timing signal, “RefCK”, thatexhibits a timing event per bit interval of a transmitted or receivedwaveform. Because each bit interval spans a respective set of N timingevents within the over-clocked timing reference, an i^(th) one of thosetiming events may be selected to time signal reception or transmission,with selection index ‘i’ ranging from 0 to N-1 and adjustable via timingcalibration executed at system startup and occasionally thereafter. Forexample, in a 16× over-clocked timing reference (16 timing events perreceived or transmitted bit), every 7^(th) timing event may initially beselected for signal reception (or transmission) timing and adjusted upor down in periodic (or occasional) timing calibration thereafter tocompensate for timing drift (e.g., progressing to the 8^(th) timingevent and then to the 9^(th,) or back to the 7^(th), etc.).

In a number of plesiochronous embodiments, the over-clocked timingreference may have an arbitrary frequency relation to an independentlygenerated data-rate timing signal (also referred to herein as RefCK)such that a non-integer number of over-clocked timing cycles transpireper bit time, t_(bit). In such embodiments, the over-clocked timingreference may be reset (or re-timed) and corresponding event-timingcounter may be reset during each cycle of the data-rate timing signal toimplement a per-bit sample-timing or transmit-timing vernier. As inmesochronous embodiments, a particular one of the multiple over-clockedtiming events that transpire in each bit time may be selected to timesignal reception and/or transmission.

FIG. 1 illustrates an exemplary signaling system 100 in whichconstituent integrated circuit devices (IC's) 101 and 131 signal oneanother from disparate temperature domains and in which an overclockedtiming reference is used to enable purely digital signal timingadjustment. In the embodiment shown, both temperature domains are lowtemperature domains, but the colder “superconducting” domain is cooledbelow the critical temperature of some or all on-chip conductors (e.g.,4 Kelvin or “4 K”) to enable implementation of exceedingly fast and lowpower circuits based on superconducting “Josephson-junction” (JJ)circuit elements. The “warmer” domain temperature is shown as 77 K andfunctional circuit blocks within the resident IC 131 (referred to hereinas the “warm IC” in contradistinction to “cold IC” 101) may beimplemented by resistor-transistor logic (RTL) and thus by complementarymetal oxide semiconductor (CMOS) circuit elements or the like. Whilethese exemplary temperature domains and circuit element implementations(i.e., JJ-based circuits in the superconducting domain, RTL circuits inthe warm domain) are carried forward in various embodiments below, inall cases the exact temperatures of the colder and warmer domains may behigher or lower than those shown, and the circuit elementimplementations may vary.

Still referring to FIG. 1, the cold and warm ICs (101 and 131) areinterconnected by a number of signaling links 108, 109 that permitbidirectional and/or unidirectional transmission of data signals,command signals, address signals, control signals, timing signals or anyother signals as necessary/useful to coordinate operations of the systemor subsystem. As a point of terminology, unless otherwise made clear bycontext, reference herein to “data” or “data signal” should beunderstood to encompass any and all information-bearing signals and“timing signal” should be understood to be a clock signal, strobe signalor any other signal that conveys timing information between the subjectICs. Moreover, while convenient in many instances to refer to “clocksignals,” and “clocking” in connection with signal-transfer timing, inall such cases, strobe signals or other occasionally idled (paused)timing signals may be used instead of continuously oscillating clocksignals.

Although two oppositely-directed unidirectional differential signalinglinks 108 and 109 are depicted in FIG. 1, any number of signaling linksmay be implemented in alternative embodiments, and any one or more ofsuch signaling links may be bidirectional and/or single-ended. Also,while virtually any chip-to-chip signaling system may be implementedusing the techniques and circuit arrangements disclosed herein (e.g.,wherever signal input/output and/or network interface is useful), forpurposes of example, the signaling system of FIG. 1 and embodimentsdescribed below are assumed to constitute a memory system or portionthereof in which a memory control function is implemented within cold IC101 (i.e., within the superconducting-domain) and a memory function(data storage) is implemented within warm IC 131. In such example, coldIC 101 may be a CPU (having one or multiple CPU cores), a dedicatedmemory control component, or any other device having a memory controlfunction, and warm IC 131 may be a memory IC (e.g., having a corestorage array implemented by dynamic random access memory, static randomaccess memory, NAND and/or NOR Flash memory or any other type of on-chipnonvolatile or nonvolatile storage cell array). In alternativeembodiments, physical signaling interfaces as shown in FIG. 1 may beused for any high-speed link or links (i.e., networking, storagecontrollers, IO Hubs, PCIe, etc.), particularly those that conveysignals between superconducting (“cold”) and cryogenic (“warm”)temperature domains.

In the exemplary embodiment of FIG. 1, signaling between the cold andwarm ICs is carried out within counterpart synchronous physicalsignaling interfaces 103 and 133 (PHYs) at a data rate established bythe aforementioned reference clock signal, RefCK. As discussed below,the RefCK frequency may be increased as timing phase alignment becomesmore precise, with higher RefCK frequencies corresponding to higherper-link data rates and thus higher system signaling bandwidth.Respective RefCK instances within the warmer and colder domains may bemesochronous (i.e., derived from a common frequency source and thusfrequency matched despite arbitrary and time-varying phase offset) orplesiochronous (generated by separate frequency sources ensured toexhibit frequency inequality no greater than a specifieddeviation/tolerance). Moreover, even in mesochronous environments, oneor more overclocked timing references within the superconducting domainmay be plesiochronous with respect to RefCK, though re-timed withrespect to RefCK timing events in at least one embodiment.

Referring specifically to warm IC 131, outbound data is transferred fromIC core 132 (e.g., memory core and access circuitry within a memory ICor data routing circuitry within a buffer IC) to data transmit circuitrywithin PHY 133. In the particular embodiment shown, for example, theoutbound data passes through multiplexer 155 (which selectively enablesa calibration data loopback as discussed below) and clocked intotransmit register 153 in response to edges (transitions) of a clocksignal, CK. The “timing” edges of CK may be limited to exclusivelyrising or exclusively falling edges, or may include both rising orfalling edges, but in either case clock a stream of outbound data bitsinto transmit register nominally at the RefCK rate to enable signaldriver 151 to drive a corresponding time-varying low-voltage-swingdifferential signal (LSDiff) onto signaling link 108. Ideally, each bitwithin the transmitted bit stream is valid on the signaling link (andthus at the cold-IC PHY) for a respective “t_(bit)” interval (i.e.,respective RefCK period) such that the transmitted bit may be sampledwithin the cold IC PHY (103) at any point in the t_(bit) interval. Inreality, time is required to effect a voltage swing on the signalinglink and at the cold IC PHY input so that the viable sampling intervalfor the transmitted bit (i.e., the data eye width) is somewhat brieferthan t_(bit). Moreover, various sources of signaling noise andsetup/hold time constraints within the cold IC PHY further constrain thetiming window for sampling (capturing) a given data bit with desiredreliability, so that ability to precisely align and maintain alignmentof the data receive sampling instant directly impacts the minimalt_(bit) duration and thus the maximum signaling rate between the twoICs.

Still referring to warm IC PHY 133, data inbound via differential link109 is converted from LSDiff to CMOS signaling levels withinamplifier/level-converter 141 and then clocked into register 143 inresponse to edges of a quadrature clock signal—that is, a clock signal(CK+ϕ) phase-shifted relative to transmit clock CK by an angle, ϕ,nominally equal to t_(bit)/2, such that, in an ideal alignment, CK+ϕtriggers sampling (within register 143) of each bit of the inbound datastream at the nominal center of the data eye.

Data flow in cold IC 101 is reversed relative to that of warm IC 131,with inbound data from link 108 received via converter/amplifier 111 andclocked into receive register 113 for delivery to the cold IC core 102.Conversely, transmit data from IC core 102 is clocked into transmitregister 123 and output via signal driver 121 onto link 109. In theparticular embodiment shown and others described below, the cold IC PHY103 is implemented with rapid single-flux-quantum (RSFQ) logic elementsor like technologies in which Josephson junctions are used to controlthe transfer of short-lived low-amplitude quantized pulses betweensuperconducting storage elements and thereby enable implementation ofextremely fast, low-power digital circuits. Various embodiments of thesedigital circuits are detailed below.

Still referring to FIG. 1, the finite signal propagation time oversignaling links 108 and 109 imposes an arbitrary(propagation-time-dependent) phase offset between counterpart clocksignals (or strobe signals) used to time data reception and transmissionin the cold and warm ICs—a phase offset initially unknown at systemstartup and subject to drift over time due to changes in voltage andtemperature. While precision analog timing calibration circuits may bedeployed within warm IC 131 to provide timing compensation (i.e.,account for the time-varying phase offset), such circuits tend todissipate considerable power; an expensive proposition even in therelatively warm (e.g., 77 K) chamber as a multiple of the timingcalibration power is typically required to extract the resulting heatenergy. Matters are complicated further in cold IC 101where variousanalog timing circuit techniques that form the backbone of timingalignment circuits in modern high-speed chip-to-chip signalingtechnologies (e.g., phase mixing circuits and phase interpolators) arerendered impracticable by the quantized nature of Josephson-junctionsignal transfer.

In the embodiment of FIG. 1, high-precision timing phase alignment isachieved with exceedingly low power expenditure through implementationof digital phase adjustment circuitry exclusively within thesuperconducting domain. More specifically, cold IC 101 includes withinPHY 103 a digital receive-phase adjuster circuit 115 that extractsselected timing events (“phases”) within an overclocked timing referenceto phase-align a receive clock signal CK_(R) with the data-eye midpointsof the incoming bit stream. In the asymmetric calibration scheme shown(i.e., calibration of both the transmit and receive signaling directionsimplemented entirely in one IC), cold-IC PHY 103 additionally includes adigital transmit-phase adjuster 125 to extract selected phases of theoverclocked timing reference and thereby advance or delay datatransmission as necessary to center (or otherwise align) data-eyeswithin a transmitted bit stream with respect to the warm-IC receiveclock (CK+ϕ). Note that the expression “data-eye,” often used inconnection with NRZ (non-return-to-zero) signaling waveforms (i.e.,logic bit conveyed as an “eye” that opens at the start of the data-validwindow and remains open until the close of the data-valid window), isused herein to refer also to the available sampling interval forreturn-to-zero (RZ) waveforms. In the case of quantized pulse streamsgenerated within PHY 103, for instance (an example of RZ signaling), the“data eye” can be viewed as the time window in which signal receiver 111may emit a quantized pulse. In either case, NRZ or RZ, timing alignmentrefers to adjustment of the signal sampling instant to a desiredtemporal position within the data-valid window or data eye.

As discussed in greater detail below, one or more error detectingcircuits are also provided within the cold-IC PHY 103 (or core circuitry102) to enable identification of upper and lower boundary phases at orbeyond which received bit error rate (BER) exceeds a hard-wired orprogrammed failure threshold. In the embodiment of FIG. 1 these errordetect circuits are shown at 117 and 127 (within phase adjusters 115 and125, respectively), with receive error detector 117 outputting a receivephase adjustment signal (Adj_(R)) to receive phase controller 119 toestablish the receive clock phase (i.e., phase of CK_(R)) midway betweenthe boundary phases identified during cold-side receive timingcalibration, and transmit error detector similarly outputting a transmitphase adjustment signal (Adj_(T)) to transmit phase controller 129 toestablish the transmit clock phase (phase of CK_(T)) midway betweenboundary phases identified during cold-side transmit timing calibration.In the particular implementation shown, each of phase controllers 119and 129 receives the data rate reference clock (RefCK)—a timingreference which may be generated within either cold IC 101 or warm IC131 (and forwarded from one IC to the other) or within anotherintegrated circuit device.

FIGS. 2A and 2B illustrate exemplary receive and transmit timingcalibration operations that may be carried out to identify receive andtransmit boundary clock phases, respectively (and center or otherwiseestablish CK_(R) and CK_(T) therebetween). Referring first to FIG. 2A, apredetermined calibration data pattern is transmitted by the warm IC andsampled within the cold IC receiver PHY in response to timing edgeswithin a leading-edge receive-boundary clock CK_(RY). The CK_(RY) clockphase is incrementally delayed from a minimally-delayed (most-advanced)timing point until samples of the incoming calibration data patternmatch expected values (i.e., BER below threshold error rate) and thenincrementally advanced to confirm the failure boundary at the leadingedge of the data eye in each sampled calibration bit. Thereafter, thedigital receive-phase adjuster begins sampling the incoming calibrationdata pattern in response to timing edges within trailing-edgereceive-boundary clock CK_(RZ), incrementally increasing the delay inthat boundary clock until the failure boundary is identified at thetrailing edge of incoming data eyes (i.e., delaying until thecalibration data is successfully received—BER below threshold—and thencontinuing to increment the delay until the BER again rises above thefailure threshold). Having identified the leading-edge and trailing-edgereceive timing boundaries, the receive-phase adjuster positions therun-time receive clock, CK_(R0), at the nominal midpoint (exact midpointor with programmed offset from exact midpoint) between the failureboundaries and thus at the desired sampling point with respect to datatransmitted by the cold IC. FIG. 2A illustrates this arrangementpost-calibration, showing edge-alignment between the warm-IC transmitclock (CK) and bitstream (DQ_(warm)), an instance of the transmittedbitstream after propagating across the signaling link (DQ_(cold)),leading-edge and trailing-edge receive timing boundaries (or “failure”boundaries) marked by CK_(RY) and CK_(RZ), and CK_(R0) centered betweenthose boundaries. FIG. 2B illustrates a post-calibration view of thecold-side transmit timing calibration. In this case, the CK_(TY) andCK_(TZ) phases that mark failure boundaries in the warm-side receiver(e.g., as determined by circulating the warm-side calibration samplesback to the cold-side receiver via the loopback path through multiplexer155 of FIG. 1) and corresponding warm-side data alignments with respectto CK+ϕ. As shown, CK_(T0) is centered between the failure boundaries sothat, after the signal propagation delay, data transmitted by thecold-side PHY arrives within the warm-side PHY in the desired quadraturealignment with the CK+ϕ sampling clock.

FIG. 3 illustrates a more detailed embodiment of asuperconducting-domain PHY 180 having RSFQ-implemented receive andtransmit timing calibration circuitry. In the inbound data path, alow-voltage-swing differential signal is received withinconverter/amplifier 187 and output as an RSFQ-level pulse stream toregisters 189 ₁-189 ₃ (referred to generally or collectively asregisters 189—a convention applied herein with respect to like-numberedelements distinguished by different subscripts), with registers 189daisy-chained to form a shift register clocked by receive clock, CK_(R).As shown, each of the register outputs is supplied to a “cycle”multiplexer 191 which responds to a cycle-select signal (CySel) byselecting the output of one of the three registers 189 as the receiveddata bit (RxData), thereby providing coarse timing adjustment over arange of three whole CK_(R) cycles. More or fewer daisy-chainedregisters 189 may be provided in alternative embodiments to expand orreduce this whole-cycle timing adjustment capability.

Still referring to FIG. 3, each data bit passed by multiplexer 191 issupplied, together with receiver-calibration pattern data from acalibration pattern data source 211 (and optionally delayed by delayelement 213 to achieve nominal alignment), to pattern compare circuit215. When enabled by a receive-calibration control value, Cal_(R),compares the received data stream with the receiver-calibration patterndata, outputting a receive-timing adjust signal (Adj_(R)) in respectivestates to indicate data match or mismatch. Though not specificallyshown, pattern compare circuit 215 may determine a bit error rate fromthe match/mismatch results (i.e., indicative of the rate at which bitsin the received data stream fail to match the expected data pattern) andraise or lower Adj_(R) according to whether the bit error rate (BER)exceeds a threshold error rate. In any case, Adj_(R) is supplied to areceive phase control circuit 181 to enable phase adjustment of thereceive boundary clocks and the run-time receive clock, for example, asdiscussed in reference to FIGS. 1 and 2A. In depicted phase controllerembodiment, a receive clock multiplexer 227 selects one of three receiveclock generators 221, 222 or 223—including respective clock generatorsfor the leading and trailing boundary clocks, CK_(RY) and CK_(RZ), and aclock generator for the run-time receive clock CK_(R0)—to output atiming control tuple that includes the CySel signal supplied tocycle-select multiplexer 191 (i.e., to enable whole-cycle receive timingadjustment) and the receive clock signal (CK_(R)) supplied to the datasampling registers 189.

The data transmit path within RSFQ PHY 180 is essentially the reverse ofthe data receive path, with normal-path transmit data (TxData) passingthrough data multiplexer 201 and then propagating through daisy-chainedregisters 203 (i.e., shift register formed by register 203 ₁, 203 ₂ and203 ₃) to enable whole-cycle timing adjustment via cycle multiplexer205. As shown, cycle multiplexer 205 supplies the cycle-selectedtransmit data value to signal driver 207 for level conversion (i.e.,from quantized RSFQ pulse to low-voltage-swing differential signalinglevel) and output onto an external signaling link.

Data transmission timing is controlled by transmit phase controller 183which outputs a time-varying transmit clock signal (CKT) to registers203 and CySel signal to cycle multiplexer 205. During transmit timingcalibration—signaled by a transmit-calibration control value,Cal_(T)—calibration pattern data from pattern generator/source 211 isselected via multiplexer 201 (i.e., instead of normal-path data from theIC core) and transmitted in accordance with the CK_(T) phase andwhole-cycle offset. The transmitted calibration data is looped back tothe data receiver within RSFQ PHY 180 via the recipient IC PHY (e.g., asshown in FIG. 1), thus enabling timing error detector 185 to generate atransmit-timing adjust signal (Adj_(T)) indicative of calibrationpattern match/mismatch (note that delay element 213 may be switched intothe calibration pattern data path during transmit timing calibration todeliver the calibration data pattern to pattern compare circuit with adelay nominally equal to the loopback path delay). As withreceive-timing calibration, Adj_(T) is applied within transmit phasecontroller 183 to adjust the phases of the transmit boundary clocks,CK_(T)y and CK_(TZ) (i.e., to identify failure timing boundaries asdiscussed above), and center the run-time transmit clock, CK_(T0),between the failure boundaries. Transmit clock multiplexer 237 selectsone of three transmit clock generators 231, 233, 235 (i.e., for CK_(TY),CK_(TZ) and CK_(T0), respectively) to output the CySel and CK_(T) tupleduring calibration operations (CK_(TY) and CK_(TZ), one after the otheras shown, for example, in FIG. 2B) and for live data transmission(CK_(T0)).

In the embodiment of FIG. 3 and others discussed below, clock generators221-225 and 231-235 are identically implemented and each receive anover-clocked timing reference, referred to herein as “fast clock” orFCLK. In a number of implementations the fast clock frequency is atleast eight times greater than the RefCK frequency (i.e., “8×”overclocking factor), and may be as high as 100× or more. In otherembodiments the overclocking factor may be as low as 4×, though, as willbecome clear, higher overclocking factors provide greater timing controlresolution and thus smaller quantized phase error. In the mesochronousimplementation of FIG. 3, FCLK is generated by on-die clock generatingcircuit 200 and frequency-divided to yield RefCK such that the FCLKfrequency is an integer multiple of the reference clock frequency (e.g.,a power-of-two multiple where divide-by-two clock dividers are used).RefCK may also be forwarded (transmitted) to the warm IC to serve as (orenable derivation of) the CK and CK+ϕ timing signals discussed above. Inother plesiochronous embodiments, discussed in greater detail withrespect to FIGS. 13-18, the fast clock and RefCK signals may begenerated independently so that the fast clock frequency may be anon-integer (and time-varying) multiple of the RefCK frequency.

FIG. 4 illustrates an exemplary phase adjustment/selection operationthat may be carried out within any one of the receive clock generators221-225 of FIG. 3 to generate respective receive clock signals CK_(RY),CK_(RZ) or CK_(R0) (corresponding operations may be executed withinclock generators 231-235 to generate transmit clock signals). In theparticular implementation shown, the over-clocked timing reference (fastclock, FCLK) oscillates at 100 GHz (10 picosecond interval betweenquantized timing pulses) and is subdivided by 16 (e.g., by a sequence offour binary frequency dividers not specifically shown) to generate a6.25 GHz RefCK signal. As discussed above, the RefCK signal correspondsto the inbound and outbound signaling rate so that the 160 picosecondRefCK period corresponds to the nominal bit-valid interval (t_(bit)) ofa symbol at any point along the serial data path, on- or off-chip. Asfurther explained above, the desired sampling instant of an inbound datasignal and the desired transmit instant of an outgoing data signal mayhave an arbitrary phase with respect to the RefCK instance within thecold-IC PHY. Accordingly, in the embodiment of FIGS. 3 and 4, phaseselect circuitry 240 within each of clock generators 221-225 enablesadjustable selection of any one of the sixteen FCLK timing pulses thatoccur per RefCK period to be output as the subject clock signal (e.g.,CK_(RY), CK_(RZ) or CK_(R0) in the case of the receive clockgenerators), thus providing a 10 picosecond timing adjust resolution(6.25% of t_(bit) per phase step). As discussed below, in oneembodiment, phase selector 240 responds to the state of the incomingadjust signal, Adj_(R) (which may be a two-bit signal capable ofrepresenting up, down (‘dn’) and no-change signal states), byincrementing or decrementing the FCLK timing phase selection and thusretarding or advancing the receive clock phase. In the example shown,CK_(Rn) (representative of CK_(RY), CK_(RZ) or CK_(R0)) is incrementallyadjusted to select FCLK pulse 9 (from the sixteen available timingpulses 0 to F, hexadecimal) and thus establish a 90 picosecond phaseoffset relative to RefCK (i.e., t_(PhO)=90 picoseconds). Transmit clockgenerators may carry out the same timing pulse selection in response toAdj_(T) to calibrate the phases of clock signals CK_(TY), CK_(TZ),CK_(T0).

FIG. 5 illustrates a more detailed embodiment of a receive clockgenerator 250 that may be used to implement any or all of receive clockgenerators 221-225 within the RSFQ PHY of FIG. 3 (which same generalcircuit architecture may be used to implement the transmit clockgenerators). As shown, clock generator 250 includes a phase counter 253that increments a modulo phase count (“PhCnt”) in response to FCLKtiming pulses (thus counting the timing pulses), and a phase selectcircuit 251 that generates an output timing pulse within output clockCK_(Rn) each time the phase counter advances to a selected phase countvalue. Phase select circuit 251 (or “phase selector”) includes aphase-adjust logic circuit 255 that, when enabled by thereceive-calibration control value Cal_(Rn), responds to the state of thereceive-timing adjust signal (Adj_(R)) by asserting count-up andcount-down signals “up” and “dn” (or deasserting both signals if noadjustment is signaled) to increment and decrement a clock-selectcounter 257. In the embodiment shown, the least significant bits of theclock-select count are supplied as a “count-match” value to comparator259. Comparator 259, accordingly, generates a timing pulse on theCK_(Rn) output each time the phase count advances to (and thus matches)the count-match value. In the 16× over-clocking example of FIG. 4, forinstance, a count-match value (lower four bits of the clock-selectcount) of nine (9) will generate a CK_(Ra) pulse at every sixteenthcycle of the FCLK signal that increments the modulo-16 phase counter to9.

Still referring to FIG. 5, the most significant bits of the clock-selectcount represent whole t_(bit) intervals and thus may be output as thecycle-select value (CySel) discussed above. That is, as the count-matchvalue overflows from 15 to 0 or underflows from 0 to 15, the cycleselect value is incremented or decremented, respectively, to select theoutput of a different one of daisy-chained receive registers (e.g., 189of FIG. 3) via the cycle multiplexer discussed above. The same operationmay be carried out to effect whole-cycle timing adjustments within thedata transmit path.

FIG. 6 illustrates embodiments of a self-timed modulo-16 phase counter281 and self-timed count-comparator 283 (e.g., that may implement thephase counter and comparator shown in FIG. 5), and also a self-timedcount-match register 285 that may store a 4-bit count-match valuegenerated, for example, by the clock-select counter of FIG. 5. In theimplementation shown, logic elements within phase counter 281 areimplemented by RSFQ circuit elements that operate analogously to RTLcircuit elements except that each of such elements require a quantumtiming pulse input to trigger a digital output (i.e., logic ‘1’indicated by presence of quantum output pulse, or logic ‘0’ indicated byabsence of quantum output pulse). FIGS. 7A-7I and FIG. 8 detail thisoperation with respect to various RSFQ circuit elements shown in FIG. 6and others deployed in embodiments described below. More specifically,each of FIGS. 7A-7I illustrates the RTL model for a given RSFQ circuitelement including:

-   -   a buffer element 331 (FIG. 7A) that generates a quantum pulse        output in response to a quantum pulse input.    -   a splitter element 341 (FIG. 7B) that generates matching quantum        pulse outputs in response to a quantum pulse input (splitter        elements may be coupled in a tree structure to yield an        arbitrary number of same-phase quantum pulse outputs and/or        staggered-phase quantum pulse outputs at various leaf nodes).    -   a join element 351 (FIG. 7C) which merges temporally proximal        quantum pulse inputs into a single quantum pulse output.    -   a register element 361 (FIG. 7D) which latches an incoming        quantum pulse, if any, and generates an output according to the        latched state (i.e., quantum pulse out if latched input pulse,        no quantum pulse out if no input pulse latched) when triggered        by a timing pulse (T). As can be seen in the RTL model, the        register element may be viewed as an asynchronous SR latch that        is set in response to a logic-true input and destructively read        out in response to the timing pulse (i.e., the timing pulse        resets the state of the RSFQ register element).    -   a data-register element 371 (FIG. 7E) which operates in the same        manner as the register element of FIG. 7D, but additionally        includes a non-destructive readout port (Q2) responsive to a        timing pulse (S).    -   a toggle-register element 381 (FIG. 7F) which toggles between        true and complement states in response to a timing pulse (T),        outputting a quantum pulse via one output port (Q+) upon        transitioning to the true state and outputting a quantum pulse        via the other output port (Q−) upon transitioning to the        complement state.    -   an XOR element 391 (FIG. 7G) which generates a quantum output        pulse in response to timing pulse (T) according to the        exclusive-OR of the state of its inputs during the interval        since the prior timing pulse assertion. That is, if, during the        interval since the prior timing pulse assertion, a quantum pulse        has been received on either of the inputs and not on the other,        the XOR element will output a quantum pulse. As in the case of        the RSFQ register element and data-register element, the readout        operation (triggered by timing pulse, T) clears the state of the        RSFQ XOR element (modeled by clearance of the two SR latches        shown in the RTL analog), thus readying the XOR element for a        subsequent XOR operation.    -   an OR element 401 (FIG. 7H) which operates generally as        described with respect to the XOR element of FIG. 7G, but        generating—in response to timing pulse, T—a logic OR of the        captured input states.    -   an AND element 411 (FIG. 7i) which also operates generally as        described with respect to the XOR element of FIG. 7G, but        generating a logic AND of the captured input states in response        to the timing pulse (T).

FIG. 8 illustrates an exemplary Josephson-junction implementation 421 ofthe RSFQ register element 361 of FIG. 7D and signal timing therefor. Thesymbol and RTL model of the register element are shown at 423 and 425,respectively. As shown, an incoming quantum pulse (φ₀ having voltagestep ΔV and width Δt) at input node “IN” transfers sufficient energyacross Josephson-junction J₂ to flip the direction of a current (I_(L))infinitely circulating (superconducted) through Josephson junctions J3and J4 and inductance L, thereby latching a “set” logic state within theregister element. Thereafter, a quantum pulse (T) delivered at node T,selectively enables either a quantum pulse output at output node “OUT”or no quantum pulse at the output node according to whether the registerelement has been set by an input quantum pulse. Exemplary relationshipsbetween a bias current I_(B) and critical current I_(C) (and componentsthereof flowing in inductive elements L₁, L₂ and L₃) are shown at 427,together with exemplary inductance values and quantum pulsetime-amplitude values. Characteristic setup and output-delay times(t_(SET) and t_(OUT), respectively) are shown in an exemplaryinput/output timing diagram at 429.

Returning now to FIG. 6, it can be appreciated that the logic outputfrom each RSFQ register element and logic gate (XOR and AND) withinmodulo phase counter 281 and from each logic gate (XOR and AND) withincomparator 283 is triggered by a timing pulse, and further thatpropagation of a logical output through a canonical coupling (i.e.,output of one RSFQ element coupled to input of another) of two or moreRSFQ circuit elements requires a temporal offset between the timingpulses delivered to those logic elements sufficient to account foroutput pulse generation time (t_(OUT)) from the upstream element andsetup time (t_(SET)) for the downstream element. Accordingly, in theembodiment of FIG. 6 and other RSFQ embodiments described below, aself-timing pulse generator 287 is provided to generate a time-staggeredset of timing pulses (depicted as T1, T2, T3, . . . , Tn) in response toeach FCLK pulse. As shown, T1 clocks a phase count value out ofcounter-state register elements 291 ₀-291 ₃ (i.e., a set of four RSFQregister elements), with each bit of the count value being replicated inmultiple instances via a respective splitter element 295 ₀-295 ₃. Ingeneral, the splitter delay is less than the time-stagger intervalbetween successive self-timing pulses (i.e., less than t_(stagger) asshown with respect to self-timing pulse generator 287) so that the phasecount output appears at the input of the modulo counter logic gates (ANDgates 301, 303 and 305 and XOR gates 297 ₀-297 ₃ shown within counter281) before delivery of timing pulses T2, T3 and T4. Accordingly, theleast significant bit of the phase count (output from register element291 ₀ and replicated in splitter 295 ₀) arrives at and is capturedwithin XOR gate 297 ₀ prior to assertion of timing pulse T3 at the otherinput of the XOR gate, and the T3 pulse occurs before timing pulse T4enables the XOR output. The XOR output (i.e., from gate 297 ₀) isfeedback to the input of counter-state register 291 ₀ and will itself bea logic ‘1’ (quantum pulse) if the least significant bit of the phasecount is ‘0’ (i.e., no quantum pulse output from register ## in responseto timing pulse T1) and a logic ‘0’ (no quantum pulse) if the leastsignificant bit of the phase count is ‘1’ and thus will cause the leastsignificant phase count bit output from register 291 ₀ to toggle atevery FCLK cycle. Applying this same analysis to the counter slices forremaining phase count bits 1, 2 and 3, it can be seen that phase countbit 1 (captured within register 291 ₁) will change state every otherFCLK cycle, phase count bit 2 (within register 291 ₂) will change stateevery fourth FCLK cycle, and phase count bit 3 (within register 291 ₃)will change state every eighth FCLK cycle, thus implementing a modulo 16counter within the four counter slices.

XOR gates 307 ₀-307 ₃ within comparator 283 each receive a respectivebit of an active-low count-match value (i.e., “CountMatch[0]−” signifiesa logic ‘0’ by delivering a quantum pulse and a logic ‘1’ by absence ofa quantum pulse) at one input and a respective active-high phase countbit at the other. Accordingly, when (and only when) the count-matchvalue matches the phase-count value, quantized pulses (logic ‘1’) willbe generated by all four XOR gates in response to timing pulse T2, whichwill in turn yield logic ‘1’ outputs from AND gates 309 and 310 inresponse to timing pulse T3 and finally a logic ‘1’ output from AND gate311 in response to timing pulse T4. Thus, in the modulo-16 counterarrangement shown, comparator 283 will generate, as CK_(Rn), a quantizedoutput pulse once every 16 FCLK cycles—once every tbit interval—at aphase offset in accordance with the match-count value. Accordingly, thematch-count value may be increased or decreased to retard or advance thephase of the CK_(Rn) timing signal, with 10 picosecond phase-adjustresolution.

In the embodiment of FIG. 3 the self-timed count-match register 285(which may constitute at least part of the counter state within theclock-select counter of FIG. 5), is implemented by data-registerelements 319 ₀-319 ₃ that may be non-destructively read-out in responseto timing pulse T1 (thus yielding active-low count-count match outputsto the inputs of comparator XOR gates 307 in time for the T2-triggeredXOR operation). As shown, the individual count-match register elements319 may be loaded with an updated count value (CountMatch′[3:0]−) inresponse to any pulse after T1 (illustrated as ‘Tn’).

Referring again to the phase counter implementation in FIG. 3, becausethe phase count value is clocked out of phase-count registers 291 atevery timing pulse T1 (i.e., once per FCLK cycle), the output phasecount triggered by a given T1 pulse must propagate through thecount-increment logic gates (XOR gates and AND gates) to produce andload the next count value back into those same phase-count registersbefore the next T1 timing pulse fires. That is, the minimum timerequired to return the next-count value to the phase-count registers—atime established by the longest chain of gates in the feedback pathbetween a phase-count register's output and input and thus in themost-significant-bit counter slice—constrains the maximum practicable(ceiling) FCLK frequency. The FCLK ceiling frequency, in turn,establishes the phase-adjust resolution (over-clocking factor) for agiven data rate and, conversely, the data rate ceiling for a givenphase-adjust resolution. In the parallel-count-load implementation shown(i.e., all counter state registers 291 loaded in parallel), four timingpulses are needed to deliver an updated state value to themost-significant count-bit register—the critical timing path. Assuming at_(stagger) in the neighborhood of 2.5-3.0 picoseconds and a 16×over-clocking factor, the FCLK ceiling frequency becomes ˜100GHz (i.e.,to provide sufficient time for timing pulses T1-T4 to deliver theupdated phase-count value to the counter-state register 291 ₃) and thedata-rate ceiling becomes 6.25 Gb/s per link.

FIGS. 9 and 10 illustrate alternative phase counter embodiments havingshortened critical feedback paths between phase-count register outputand input to permit higher FCLK frequencies and correspondingly highersignaling rates (or increased phase-adjust resolution for a givensignaling rate). In the exemplary ripple-counter 450 of FIG. 9, aphase-count value is output from each of N count-state registers 451₀-451 ₁ in sequence, starting with the least significant bit andprogressing bit by bit to the most significant bit. By this arrangement,the carry-bit supplied to a given count-slice from its less significantneighbor may be generated (within gate 459 of each counter slice) inparallel with the count-register output, thus avoiding the accumulatedBoolean operation delays that yield the 4-gate (4 timing pulse) criticalpath in the parallel counter implementation of FIG. 6. Morespecifically, the critical path within each count slice (i.e., fromcount register output through one of splitters (S) and XOR gate 461 backto count register input) requires merely two timing pulses—half thatrequired in the parallel counter implementation of FIG. 6. Accordingly,the FCLK ceiling is doubled from ˜100GHz to ˜200GHz (i.e., assuming thesame exemplary t_(stagger) and 16× over-clocking factor discussed inreference to FIG. 6), and the data rate is correspondingly doubled from6.25 Gb/s/link to 12.5 Gb/s/link.

Still referring to FIG. 9, it should be noted that the accumulatedripple delay across the counter is of no consequence to the FCLK anddata rate ceilings as canonical operation within the counter (i.e.,propagation of logic signals canonically as opposed to being fed back)may simply be pipelined with final output of the counter and/orcomparator occurring one or more FCLK cycles after the FCLK cycle thatyields a given phase-count output.

FIG. 10 illustrates an exemplary toggle counter 470 in whichcounter-state feedback is implemented within the counter-state registerelements themselves (i.e., the count register within each counter slice471 ₀-471 _(n−1) is implemented by a toggle register 473 that changesstate in response to each input timing pulse), thus permitting thecounter to increment from count to count with virtually no delay savethat required for the toggle element to change state. Such embodimentsmay execute at, near or above terahertz FCLK frequencies. In onetoggle-counter embodiment, for example, an 800 GHz FCLK with 16×overclocking yields a 50 Gb/s data rate with 1.25 picosecondphase-adjust resolution.

FIG. 11 illustrates exemplary per-bit count-match circuitry 490 thatgenerates a set of match bits for the toggle counter slices 471 shown inFIG. 10. In the depicted embodiment, each count-match slice 491 ₀-401_(n−1) includes a pair of data registers 493, 495 that are loaded withactive high and active low (complementary or differential) instances ofthe count-match bit for the subject counter slice. Complementaryinstances of the phase-count output by each counter slice are suppliedto trigger non-destructive read-outs from the data registers with thoseread-outs merged at join element 497 to form the match-bit output forthe corresponding count slice (i.e., element 471 of FIG. 10). By thisarrangement, if the toggle register within a given count slice outputs alogic ‘1’ count bit that matches a corresponding logic ‘1’ count-matchbit, data register 493 will deliver a logic ‘1’ output (quantized pulse)that will propagate through join element 497 to the match-bit output.Similarly, if the toggle register within a given count slice outputs alogic ‘0’ count bit (i.e., PhCnt[i]−yields a quantized pulse) thatmatches a logic ‘0’ count-match value, data register 495 will deliver alogic ‘1’ output (quantized) pulse that will propagate through joinelement 497 to the match-bit output. If the count bit and match bit donot match, neither of data registers 493, 495 will yield a logic ‘1’output, with the absence of a quantized pulse output from join element497 signaling the mismatch result.

Referring to both FIGS. 9 and 10, the logical combination of match bits(i.e., each signaling match or mismatch between a count bit andcorresponding count-match bit) to yield the output clock signal may beimplemented in a number of different ways. Regarding the ripple counterof FIG. 9, for example, the comparison of each count-match bit with arespective phase-count bit may be time-staggered in accordance with thetime-staggered (rippled) generation of the phase count bits themselves,with each match bit being logically ANDed with each subsequentlygenerated match bit to yield, after ANDing with the most-significantmatch bit, the combined match result and thus the selected clock phase.In the case of the high-speed toggle counter of FIG. 10 andcorresponding count-match circuitry of FIG. 11, the individual matchbits may be applied as respective, staggered timing pulses to propagatea logic ‘1’ seed value (e.g., chosen from a given counter state) througha high-speed shift register. By that operation, if any of the match bitsis a logic ‘0’ the shift-chain will be broken and the shift registeroutput will fail to yield a quantized clock pulse. By contrast, if allmatch bits are a logic ‘1’, the logic ‘1’ seed value will propagatethrough the shift register to yield a logic ‘1’ output (quantized pulse)each time the toggle counter advances to the count-match specified countvalue. Various other comparator circuits may be used in alternativeembodiments.

FIG. 12 illustrates an exemplary self-timing pulse generator 500 thatmay be used to generate self-timing pulses (T1, T2, . . . ) in any ofthe self-timed RSFQ logic circuit embodiments herein. As shown, T1 pulseinstances are generated by a splitter 501, with one of those T1 pulsespropagating through a wired delay path 503 ₁ (which may additionally oralternatively traverse one or more RSFQ buffer elements or other“active” delay elements in alternative embodiments) to yield a delayedT1 instance, T1d. T1d is applied in turn, to trigger output of the T1pulse captured in register 505 ₁ (resetting the register by virtue ofthe destructive read-out) and thus a timing pulse that yields a numberof T2 pulse instances via splitter 501 ₂. T3 is generated from a delayedinstance of T2 in generally the same way that T2 is generated from thedelayed instance of T1 (i.e., T2 propagates through delay line 503 ₂ toproduce T2d which, in turn, triggers output from register 505 ₂ andnumerous T3 instances via splitter 501 ₃), and T4, T5, etc., aresimilarly generated respectively from T3, T4, etc., via similar oridentical delay lines (503 ₃, 503 ₄, etc.) registers (505 ₃, 505 ₄,etc.) and splitter elements (501 ₄, 501 ₅, etc.). In a number ofembodiments, the individual delay lines (503 ₁, 503 ₂, etc.) havematched delays and are trimmed/designed to match the input setup time(with engineered margin) of the downstream register. In thoseembodiments, the pulse-to-pulse timing offset, t_(stagger), is uniformacross the asynchronous timing pulse train and cumulatively reflects thedelay-line propagation delay (t_(delay)), the output delay of thetiming-pulse-source register (e.g., t_(out) of register 505 ₁ in thecase of t_(stagger) between T1 and T2) and the propagation delay througha given splitter element (t_(split)). One or more delays may bepurposefully made larger than others in alternative embodiments to yieldnon-uniform t_(stagger) intervals between selected timing pulses.

In the mesochronous embodiments discussed thus far, the RefCK period (ort_(bit)) is an integer multiple of the FCLK period—that is, theoverclocking factor is an integer value. FIG. 13 illustrates aplesiochronous receive/transmit timing approach that avoids thisconstraint, permitting FCLK and RefCK to have an arbitrary (includingnon-integer) and even time-varying frequency ratio. In the particularexample shown, the RefCK period is ˜80 picoseconds (˜12.5 Gb/s/link datarate), while the FCLK period is on the order of ˜8.4 picoseconds suchthat nine FCLK timing pulses are nominally generated during a RefCKperiod, with a fractional interval of approximately 4.2 picoseconds leftover. If both clocks are allowed to run freely, the fractional intervalswill accumulate such that approximately once every three RefCKintervals, only eight FCLK timing pulses will be generated. In a numberof embodiments, however, FCLK is re-timed (or reset or restarted) inresponse to each RefCK timing pulse such that a nominally repeatingnumber of FCLK timing events occur per bit time. Note that the integernumber of FCLK pulses per t_(bit) interval may yet increase or decreaseover macro-time (e.g., milliseconds, seconds or longer) due totemperature/voltage-induced drift in the FCLK-RefCK frequency ratio.

Still referring to FIG. 13, by also resetting (or restarting) anFCLK-triggered phase counter in response to each RefCK timing event, arepeating count of FCLK timing events is generated during each RefCKperiod, thus providing a repeatable timing index that may be used toextract an FCLK timing pulse having a desired/calibrated phase offsetwithin each t_(bit) interval. In the particular example shown, forexample, a count-match value of ‘5’ selects an FCLK pulse having a ˜50.5picosecond phase offset within each t_(bit) interval.

FIG. 14 illustrates an embodiment of a receive clock generator 550 thatenables generation of a receive timing signal (CK_(Rn)) having adigitally controlled phase offset relative to RefCK despite aplesiochronous relationship between an over-clocked timing reference(FCLK) and RefCK. As with all receive clock generators presented herein,a transmit clock generator may be implemented generally as depicted,receiving Adj_(T) and Cal_(Tn) signals instead of thereceive-calibration control signals shown, and outputting CK_(Tn) andCySel_(Tn) instead of CK_(Rn) and CySel_(Rn).

As with the embodiment of FIG. 5, clock generator 550 includes a phaseselector 551 and phase counter 553, with constituents of phase selector551 including phase-adjust logic 561, clock-select counter 563 andcomparator 565. Phase-adjust logic 561 and comparator 565 operategenerally as discussed in reference to FIG. 5 (with comparator 565implemented by any of the variants described above). Clock-selectcounter 563 also operates generally as discussed in reference to FIG. 5with the exception that overflow and underflow from the count-matchfield (i.e., least significant N bits of the clock-select count) occurwith an arbitrary modulus established by modulus logic 554 within phasecounter 553. For example, in the exemplary FCLK/RefCK frequency ratioshown in FIG. 13, the least significant bits will be counted modulo-9for most or all RefCK cycles, overflowing from a count value of 9 to acount value of 0 (and incrementing the cycle-select value) andunderflowing from a count value of 0 to a count value of 9 (anddecrementing the cycle-select value). Modulus logic 554 serves tocapture and output this overflow/underflow modulus (‘M’) to clock-selectcounter 563, enabling drift-induced increase/decrease in the modulusvalue over time.

Still referring to FIG. 14, phase counter 553 receives the RefCK signalat a reset input and is thus reset in response to each RefCK timingpulse (i.e., once per tbit interval), an operation that may also shiftthe top-end phase count (the maximum count reached before reset) intomodulus logic 554 to establish the overflow/underflow modulus forclock-select counter 563. RefCK is also supplied to the reset input of afast-clock generator 555 and, as described in further detail below,serves to re-time (or reset or restart) the FCLK output at the start ofeach t_(bit) interval (RefCK period) such that each FCLK pulse generatedduring the t_(bit) interval exhibits a nominally fixed phase offset withrespect to the RefCK timing pulse (“nominally fixed” as there may be arelatively slow temperature and/or voltage-induced drift in the phaseoffset that can be compensated/tracked through periodic timingcalibration). As shown, FCLK is supplied to a self-timing pulsegenerator 557 (e.g., implemented generally as shown in FIG. 12) to yieldself-timing pulses to phase counter 553 and comparator 565 as necessaryto carry out clock-phase selection for each t_(bit) interval. Asdiscussed above, logic operations within phase counter 553 and/orcomparator 565 may be pipelined so that the timing pulse sequencelaunched by a given FCLK pulse may extend into a subsequent FCLK periodand possibly even multiple FCLK periods. Thus, depending on the lengthof the timing pulse chain generated within the embodiment of FIG. 12(i.e., number of pulses generated per FCLK pulse), timing pulses mayconcurrently propagate through two or more of delay lines 503.

Returning to the exemplary plesiochronous FCLK-RefCK timing relationshipshown in FIG. 13, it can be seen that an interstitial interval betweenthe final FCLK pulse within a given RefCK period and the initial FCLKpulse of the ensuing RefCK period may be arbitrarily small, tendingtoward zero in the limiting case. Absent effort to impose a minimumduration, this interstitial interval may be too brief to reset the phasecounter and/or FCLK generator in preparation for counting into the newRefCK period. And, while imposing a minimum interstitial duration toavoid such reset-timing pinch is an option in at least one embodiment,such solutions generally impose constraints on the RefCK-FCLK phaserelationship.

FIGS. 15, 16, 17 and 18 illustrate embodiments of digital phaseadjusters that avoid the reset-timing pinch by providing dual FCLKgenerators and phase counters and activating an alternate FCLKgenerator/phase counter pair from one RefCK period to the next. Morespecifically, as one FCLK generator/phase counter pair is activated toindex into a given RefCK period, the other pair is reset and made readyfor activation in the immediately ensuing RefCK period. By thisoperation, the reset operation (of both the FCLK generator and phasecounter) is removed from the critical timing path, thus avoiding thereset-timing pinch without constraining the plesiochronous RefCK-FCLKtiming relationship.

Referring to the exemplary receive clock generator shown in FIG. 15 andcorresponding timing diagram in FIG. 16, a finite state machine 601(FSM) toggles between even and odd states (StateE and StateO,respectively), transitioning from one to the other in response to eachRefCK timing pulse. In the even state, FSM 601 asserts an even-cycleenable signal (en_e) to FCLKe generator 602 to enable even-cycle FCLKgeneration at a predetermined (and repeatable) time relative to theRefCK pulse that produced the even FSM state. Also while in the evenstate, FSM 601 asserts an odd-cycle reset signal (rst_o) to odd-cyclephase counter 605 to reset the counter-state therein, thus ensuring thatthe first odd-state FCLK pulse will yield a zero-valued (or otherwisereset) phase count from the odd-cycle phase counter. As the even-cyclephase counter 604 was similarly reset upon transition from the evenstate to the odd state, the even-cycle phase counter will, upon FSMentry to the even state, count the FCLKe timing pulses from 0 to maximumthroughout the even-state RefCK period—an operation shown in FIG. 15.Thereafter, upon FSM transition to the odd state at the ensuing RefCKpulse, the FSM will deassert en_e and rst_o and instead assert en_o andrst_e, thereby enabling the FCLKo generator 603 and resetting even phasecounter 604. Accordingly, as shown in FIG. 15, the odd-cycle phasecounter will count the odd-cycle FCLK pulses (i.e., FCLKo pulses) from 0to maximum, while the even-cycle FCLK generator is idled. This cyclicaltransition between dual clock generators and phase counters continues atleast through the duration of a data burst reception for the receiveclock generator shown (or through data burst transmission in the case ofa transmit clock generator).

In the embodiment of FIG. 15, the even phase count and odd-phase countvalues are supplied to respective comparator circuits 608 and 609 forcomparison with a count-match value generated by phase selector 610,thus yielding even-cycle and odd-cycle clock pulses that are joinedwithin element 611 to yield the finalized CK_(Rn) output. As shown, thephase counter and comparator pair for a given state (even or odd) mayreceive self-timing signals from a respective even state/odd-stateself-timing pulse generator 606 and 607, with each such pulse generatorgenerating a sequence of time-staggered pulses in response to each pulseof the corresponding fast clock (FCLKe or FCLKo).

FIG. 17 illustrates a more detailed embodiment of a clock generator 625that may implement the dual receive clock generator of FIG. 16 (and thusany of the receive or transmit clock generators shown in FIG. 1). Asshown, clock generator 625 includes a finite state machine 630 (FSM)having a toggle register 635 that transitions alternately between evenand odd states (StateE′ and StateO′) during successive even and oddRefCK periods (cycles) and, for each of the even and odd states, arespective one of: FCLK generators 640 and 641 (generating FCLKe andFCLKo, respectively), self-timing pulse generators 650 and 651, phasecounters 664 and 665 and comparators 670 and 671. Upon toggle-registertransition to the even state (StateE′), a logic ‘1’ StateE signal fromsplitter 637 is latched within data register 664 of FCLKe generator 640and thereafter clocked out of the data register in response to a delayedinstance of the RefCK pulse that yielded the state transition. That is,the delayed RefCK instance from delay element 632 (RefCKd) passesthrough join element 646 and splitter 658 to yield an initial FCLKepulse. The alternate quantized pulse output of splitter 658 (i.e.,replica of the FCLKe pulse) returns to the non-destructive-readoutstrobe of data register 644 to produce, via non-destructive read-outoutput Q2, another quantized pulse that also propagates through joinelement 646 and splitter 648 to emerge as a second FCLKe pulse, withthat FCLKe pulse again triggering a non-destructive readout from dataregister 644 and thus a third FCLKe pulse. By this operation, an ongoingstream of FCLKe pulses is generated at a rate corresponding to the loopdelay (i.e., t_(out) delay of data register 644 and propagation delaysthrough join and split elements 646 and 648) until transition of FSM 630to the odd state. At that point, a StateO signal (from the odd-statecounterpart to even-state splitter 637) pulses the destructive read-outstrobe (T) of data register 644, thereby clearing the internal state ofthe data register and thus preventing further FCLKe pulse generationuntil the FSM reverts to the even state.

Still referring to FIG. 17, the even-cycle fast clock (FCLKe) issupplied to even-cycle self-timing pulse generator 650 which, in turn,generates a self-timing pulse train (T1, T2, T3, . . . Tn) in responseto each FCLKe pulse, supplying that pulse train to logic elements withinthe count slices of even-cycle phase counter 664 and to even-cyclecomparator 670. Even-cycle phase counter 664 may be implemented by anyof the various phase counter embodiments herein, with modification toenable the counter to be reset at conclusion of the counted RefCK period(i.e., upon transition to the state in which the alternate phase countertakes over). As shown in detail view 669 of an exemplary implementationof count slice 673, a data register 675 and logic AND gate 677 are addedto a ripple-count slice otherwise implemented as shown in FIG. 9 toreset count-state register 453 in preparation for counting anew during asubsequent RefCK period. Additional self-timing pulses may be generatedas necessary (i.e., after the even state terminates) to clear thecontent of register 453 and thus zero (or otherwise pre-set) theeven-cycle phase counter in preparation for the next even-cycle countingsequence. More generally, various alternative reset logic arrangementsmay be implemented in alternative embodiments (i.e., any circuitarrangement that clears or otherwise establishes a predetermined counterstate such that counting is re-started upon return to the subject FSMstate). Parallel-count slices and toggle-count slices as shown in FIGS.6 and 10, respectively, may also be supplemented with reset logic.

Still referring to FIG. 17, odd-state clock generator, self-timing pulsegenerator and phase counter operate identically as described withrespect to their even-state counterparts, but during odd RefCK cycles.Accordingly, phase counts from the even-cycle and odd-cycle phasecounters 664 and 665 are supplied to the even-cycle and odd-cyclecomparators 670 and 671, respectively, to enable selection (extraction)of even-cycle and odd-cycle receive clock pulses, CK_(Re) and CK_(Ro).In the embodiment shown, those clock pulses are joined within joinelement 675 to yield the completed (composited) receive clock, CK_(Rn).

FIG. 18 illustrates an exemplary timing diagram corresponding to FIG.17. As shown, the even-cycle fast clock (FCLKe) and odd-cycle fast clock(FCLKo) are alternately enabled along with respective even-cycle andodd-cycle phase counters to count into respective RefCK periods. Thephase counter and comparator circuitry yield a selected even-cycle andodd-cycle output clock pulse—one pulse per RefCK period with amatch-count selected phase offset—that are joined to yield the finalizedoutput clock.

FIGS. 19 and 20 correspond to an alternative receive/transmit timingadjustment technique that lack encoded phase counters and (in at leastone embodiment) phase count comparators. As shown, a recirculating shiftregister 703 (or ring buffer) iteratively rotates 1-hot pattern data(i.e., only one of the N pattern data bits is a logic ‘1’) such that asingle quantized pulse is output from the shift register once per everyN bit-shift operations. In the depicted embodiment, FCLK is supplied toshift the data pattern incrementally through the ring buffer (with FCLKpropagating through a daisy-chained set of splitter elements 701 ₀-701₁₅ to stagger the timing pulses supplied to the individual RSFQ registerelements 705 ₀-705 ₁₅ that constitute the ring buffer). As N=16 in theexample shown, one out of every 16 FCLK pulses yields a quantized pulseat the receive clock output (i.e., from splitter 707, which alsoprovides a feedback output to close the recirculation ring).

Still referring to FIGS. 19 and 20, the phasing of the receive clockwith respect to a given frame of FCLK pulses (i.e., FCLK pulses framedby a given RefCK period) may be controlled by the initial position ofthe 1-hot bit (the solitary logic ‘1’ bit) within the pattern data. Inthe particular embodiment shown, a load multiplexer 709 is provided toenable a new data pattern to be shifted into the ring buffer(overwriting the old) and thus establish the 1-hot bit at any of sixteenphase offsets within a given RefCK period. FIG. 10 illustrates anexemplary timing diagram corresponding to operation of the phaseadjuster of FIG. 19, showing a one-hot pattern that establishes thefifth FCLK pulse (i.e., pulse number ‘4’, starting from 0) within eachRefCK period as the desired receive clock phase. Note that, while nocount comparator circuitry is implemented within the embodiment of FIG.19, the N different possible 1-hot states of the ring buffer may beviewed as implementing a fully decoded counter (i.e., decoded countstates instead of encoded count states). Accordingly, instead ofestablishing a desired receive clock phase through pattern dataalignment with the RefCK frame, the decoded output could be comparedwith a decoded count-match value, or the decoded output could be encodedand compared with an encoded count-match value. In either embodiment, asthe critical timing path in the counter-state transition is limited tothe speed at which data may be shifted between any two registers 705 inwithin shift register 703, extremely high FCLK frequencies may beemployed, approaching or surpassing terahertz frequencies as in thetoggle-counter implementation discussed above. As each change in thecount-state ripples through the register elements, a correspondinglypipelined compare operation may be used to extract the desired receiveclock phase.

It should be noted that the various circuits disclosed herein may bedescribed using computer aided design tools and expressed (orrepresented), as data and/or instructions embodied in variouscomputer-readable media, in terms of their behavioral, registertransfer, logic component, transistor, layout geometries, and/or othercharacteristics. Formats of files and other objects in which suchcircuit expressions may be implemented include, but are not limited to,formats supporting behavioral languages such as C, Verilog, and VHDL,formats supporting register level description languages like RTL, andformats supporting geometry description languages such as GDSII, GDSIII,GDSIV, CIF, MEBES and any other suitable formats and languages.Computer-readable media in which such formatted data and/or instructionsmay be embodied include, but are not limited to, computer storage mediain various forms (e.g., optical, magnetic or semiconductor storagemedia, whether independently distributed in that manner, or stored “insitu” in an operating system).

When received within a computer system via one or more computer-readablemedia, such data and/or instruction-based expressions of the abovedescribed circuits can be processed by a processing entity (e.g., one ormore processors) within the computer system in conjunction withexecution of one or more other computer programs including, withoutlimitation, net-list generation programs, place and route programs andthe like, to generate a representation or image of a physicalmanifestation of such circuits. Such representation or image canthereafter be used in device fabrication, for example, by enablinggeneration of one or more masks that are used to form various componentsof the circuits in a device fabrication process.

In the foregoing description and in the accompanying drawings, specificterminology and drawing symbols have been set forth to provide athorough understanding of the disclosed embodiments. In some instances,the terminology and symbols may imply specific details that are notrequired to practice those embodiments. For example, any of the specificvoltages, pixel array sizes, signal path widths, signaling or operatingfrequencies, component circuits or devices and the like can be differentfrom those described above in alternative embodiments. Additionally,links or other interconnection between integrated circuit devices orinternal circuit elements or blocks may be shown as buses or as singlesignal lines. Each of the buses can alternatively be a single signalline, and each of the single signal lines can alternatively be buses.Signals and signaling links, however shown or described, can besingle-ended or differential. A signal driving circuit is said to“output” a signal to a signal receiving circuit when the signal drivingcircuit asserts (or de-asserts, if explicitly stated or indicated bycontext) the signal on a signal line coupled between the signal drivingand signal receiving circuits. The term “coupled” is used herein toexpress a direct connection as well as a connection through one or moreintervening circuits or structures. Integrated circuit device“programming” can include, for example and without limitation, loading acontrol value into a register or other storage circuit within theintegrated circuit device in response to a host instruction (and thuscontrolling an operational aspect of the device and/or establishing adevice configuration) or through a one-time programming operation (e.g.,blowing fuses within a configuration circuit during device production),and/or connecting one or more selected pins or other contact structuresof the device to reference voltage lines (also referred to as strapping)to establish a particular device configuration or operation aspect ofthe device. The term “light” as used to apply to radiation is notlimited to visible light, and when used to describe sensor function isintended to apply to the wavelength band or bands to which a particularpixel construction (including any corresponding filters) is sensitive.The terms “exemplary” and “embodiment” are used to express an example,not a preference or requirement. Also, the terms “may” and “can” areused interchangeably to denote optional (permissible) subject matter.The absence of either term should not be construed as meaning that agiven feature or technique is required.

Various modifications and changes can be made to the embodimentspresented herein without departing from the broader spirit and scope ofthe disclosure. For example, features or aspects of any of theembodiments can be applied in combination with any other of theembodiments or in place of counterpart features or aspects thereof.Accordingly, the specification and drawings are to be regarded in anillustrative rather than a restrictive sense.

What is claimed is:
 1. A method of operation within an integratedcircuit device, the method comprising: incrementing a modulo-N phasecount in response to respective timing events within a first timingsignal such that the phase count repeatedly sequences between N discretevalues, N being an integer greater than 4; after each phase-countincrement, comparing the phase count to a match count having a valuebetween 0 and N to produce, as a second timing signal, a match-resultsignal that indicates a match once for each repetition of the N discretevalues of the phase count; and sampling a data signal received via asignaling link external to the integrated circuit device in response tothe second timing signal.
 2. The method of claim 1 further comprisingadjusting the match count to adjust a phase-offset of the second timingsignal with respect to the data signal.
 3. The method of claim 2 furthercomprising determining at least one boundary phase-offset of the secondtiming signal that, when applied to sample the data signal, yields a biterror rate greater than a predetermined threshold, and wherein adjustingthe match count comprises adjusting the match count in accordance withthe boundary phase-offset.
 4. The method of claim 1 wherein thefrequency of the first timing signal is an integer multiple of a datarate of the data signal.
 5. The method of claim 1 wherein the frequencyof the first timing signal is a non-integer multiple of a data rate ofthe data signal.
 6. The method of claim 1 wherein N is subject to changeduring operation of the integrated circuit device due to change involtage and/or temperature.
 7. The method of claim 1 further comprisinggenerating, in response to each of the timing events within the firsttiming signal, a plurality of asynchronous timing pulses, and whereinincrementing the phase count in response to respective timing eventswithin the first timing signal comprises propagating the phase countthrough a plurality of logic elements at times controlled by theasynchronous timing pulses to generate an incremented phase count.
 8. Asystem comprising: at least one signaling link; a firstintegrated-circuit component (IC) having a first interface coupled tothe signaling link and timed by a first interface timing signal; asecond integrated-circuit component (IC) having: a second interfacecoupled to the signaling link and timed by a second interface timingsignal that is mesochronous with respect to the first interface timingsignal; and a phase adjustment circuit to adjust a phase of the secondinterface timing signal using a digital counter implemented withJosephson-junction circuit elements.
 9. The system of claim 8 whereinthe first and second ICs are disposed in disparate first and secondtemperature domains, respectively, the phase adjustment circuit beinginoperable in the first temperature domain.
 10. The system of claim 8wherein the digital counter is implemented using rapidsingle-flux-quantum (RSFQ) circuit elements having Josephson junctions.11. The system of claim 8 wherein the second IC further comprises atiming signal generator to generate an over-clocked timing signal have afrequency more than four times higher than that of the second interfacetiming signal.
 12. The system of claim 11 wherein the digital counter iscoupled to receive and count timing events within the over-clockedtiming signal.
 13. The system of claim 11 wherein the first IC is amemory control component and the second IC is a memory component. 14.The system of claim 11 further comprising one or more memory ICs andwherein the first IC is a memory control component and the second IC isa buffer component coupled between the memory control component and theone or more memory ICs.
 15. An integrated circuit device comprising: amodulo-N phase counter to increment a phase count in response torespective timing events of a first timing signal such that the phasecount repeatedly sequences between N discrete values, N being an integergreater than 4; a comparator that compares the phase count, after eachincrement thereof, to a match count having a value between 0 and N toproduce, as a second timing signal, a match-result signal that indicatesa match once for each repetition of the N discrete values of the phasecount; and sampling circuitry to sample a data signal received via asignaling link external to the integrated circuit device in response tothe second timing signal.
 16. The integrated circuit device of claim 15further comprising timing-calibration circuitry that adjusts the matchcount to change a phase-offset of the second timing signal with respectto the data signal.
 17. The integrated circuit device of claim 16wherein the timing-calibration circuitry comprises circuitry to (i)determine at least one boundary phase-offset of the second timing signalthat, when applied to sample the data signal, yields a bit error rategreater than a predetermined threshold, and (ii) adjust the match countin accordance with the boundary phase-offset.
 18. The integrated circuitdevice of claim 15 wherein the frequency of the first timing signal isan integer multiple of a data rate of the data signal.
 19. Theintegrated circuit device of claim 15 wherein the frequency of the firsttiming signal is a non-integer multiple of a data rate of the datasignal.
 20. The integrated circuit device of claim 15 wherein N issubject to change during operation of the integrated circuit device dueto change in voltage and/or temperature.
 21. The integrated circuitdevice of claim 15 further comprising self-timing pulse generatingcircuitry to generate, in response to each of the timing events withinthe first timing signal, a plurality of asynchronous timing pulses thatenable operation of the phase counter and comparator.